REV. 0

Information furnished by Analog Devices is believed to be accurate and
reliable. However, no responsibility is assumed by Analog Devices for its
use, nor for any infringements of patents or other rights of third parties
which may result from its use. No license is granted by implication or
otherwise under any patent or patent rights of Analog Devices.

ADP1110

One Technology Way, P.O. Box 9106, Norwood, MA 02062-9106, U.S.A.
Tel: 617/329-4700

World Wide Web Site: http://www.analog.com

Fax: 617/326-8703

© Analog Devices, Inc., 1996

Micropower, Step-Up/Step-Down Switching

Regulator; Adjustable and Fixed 3.3 V, 5 V, 12 V

FUNCTIONAL BLOCK DIAGRAMS

SENSE

DRIVER

LIM

SW1

SW2

SET

220mV

REFERENCE

GAIN BLOCK/
ERROR AMP

OSCILLATOR

ADP1110

GND

COMPARATOR

300k

ADP1110 Block Diagram--Fixed Output Version

DRIVER

LIM

SW1

SW2

GND

SET

220mV

REFERENCE

GAIN BLOCK/
ERROR AMP

COMPARATOR

OSCILLATOR

ADP1110

ADP1110 Block Diagram--Adjustable Output Version

FEATURES
Operates at Supply Voltages From 1.0 V to 30 V
Step-Up or Step-Down Mode
Minimal External Components Required
Low-Battery Detector
User-Adjustable Current Limiting
Fixed or Adjustable Output Voltage Versions
8-Pin DIP or SO-8 Package

APPLICATIONS
Cellular Telephones
Single-Cell to 5 V Converters
Laptop and Palmtop Computers
Pagers
Cameras
Battery Backup Supplies
Portable Instruments
Laser Diode Drivers
Hand-Held Inventory Computers

GENERAL DESCRIPTION

The ADP1110 is part of a family of step-up/step-down switch-
ing regulators that operate from an input voltage supply as little
as 1.0 V. This very low input voltage allows the ADP1110 to be
used in applications that use a single cell as the primary power
source.

The ADP1110 can be configured to operate in either step-up or
step-down mode, but for input voltages greater than 3 V, the
ADP1111 would be a more effective solution.

An auxiliary gain amplifier can serve as a low battery detector or
as a linear regulator.

The quiescent current of 300

A makes the ADP1110 useful in

remote or battery powered applications.

The 70 kHz frequency operation also allows for the use of
surface-mount external capacitors and inductors.

Battery protection circuitry limits the effect of reverse current to
safe levels at reverse voltages up to 1.6 V.

REV. 0

ADP1110SPECIFICATIONS

Parameter

Conditions

Min

Typ

Max

Units

QUIESCENT CURRENT

Switch Off

300

INPUT VOLTAGE

Step-Up Mode

1.15

12.6

Step-Down Mode

COMPARATOR TRIP POINT VOLTAGE

ADP1110

210

220

230

OUTPUT SENSE VOLTAGE

ADP1110-3.3

OUT

3.13

3.30

3.47

ADP1110-5

4.75

5.00

5.25

ADP1110-12

11.4

12.00

12.6

COMPARATOR HYSTERESIS

ADP1110

OUTPUT HYSTERESIS

ADP1110-3.3

130

ADP1110-5

180

ADP1110-12

200

400

OSCILLATOR FREQUENCY

OSC

kHz

DUTY CYCLE

Full Load (V

< V

REF

)

SWITCH ON TIME

7.5

12.5

FEEDBACK PIN BIAS CURRENT

ADP1110 V

= 0 V

150

240

SET PIN BIAS CURRENT

SET

= V

REF

SET

300

500

A0 OUTPUT LOW

= 300

0.15

0.4

SET

= 150 mV

REFERENCE LINE REGULATION

1.0 V

1.5 V

0.35

%/V

1.5 V

12 V

0.05

0.1

%/V

SWITCH SATURATION VOLTAGE

= 1.5 V, I

= 400 mA, +25

CESAT

300

500

STEP-UP MODE

MIN

to T

MAX

600

= 1.5 V, I

= 500 mA, +25

400

650

MIN

to T

MAX

750

= 5 V, I

= 1 A, +25

700

1000

A2 ERROR AMP GAIN

= 100 k

1000

5000

V/V

REVERSE BATTERY CURRENT

= +25

REV

750

CURRENT LIMIT TEMPERATURE

IN,

= +25

0.3

COEFFICIENT

SWITCH OFF LEAKAGE CURRENT

Measured at SW1 Pin,

LEAK

= +25

MAXIMUM EXCURSION BELOW GND

SW1

A, Switch Off

SW2

400

350

= +25

NOTES

This specification guarantees that both the high and low trip point of the comparator fall within the 210 mV to 230 mV range.

This specification guarantees that the output voltage of the fixed versions will always fall within the specified range. The waveform at the sense pin will exhibit a saw-

tooth shape due to the comparator hysteresis.

100 k

resistor connected between a 5 V source and the AO pin.

The ADP1110 is guaranteed to withstand continuous application of +1.6 V applied to the GND and SW2 pins while V

, I

LIM

, and SW1 pins are grounded.

All limits at temperature extremes are guaranteed via correlation using standard statistical quality control methods.

Specifications subject to change without notice.

(0 C to +70 C, V

= 1.5 V unless otherwise noted)

ADP1110

REV. 0

PIN DESCRIPTION

Mnemonic

Function

LIM

For normal conditions this pin is connected to
V

. When lower current is required, a resistor

should be connected between I

LIM

and V

Limiting the switch current to 400 mA is
achieved by connecting a 220

resistor.

Input Voltage.

SW1

Collector Node of Power Transistor. For step-
down configuration, connect to V

. For step-

up configuration, connect to an inductor/diode.

SW2

Emitter Node of Power Transistor. For step-
down configuration, connect to inductor/diode.
For step-up configuration, connect to ground.
Do not allow this pin to go more than a diode
drop below ground.

GND

Ground.

Auxiliary Gain (GB) Output. The open collec-
tor can sink 300

A. It can be left open if unused.

SET

Gain Amplifier Input. The amplifier has posi-
tive input connected to SET pin and negative
input connected to 220 mV reference. It can be
left open if unused.

FB/SENSE

On the ADP1110 (adjustable) version this pin
is connected to the comparator input. On the
ADP1110-3.3, ADP1110-5 and ADP1110-12,
the pin goes directly to the internal application
resistor that set output voltage.

ABSOLUTE MAXIMUM RATINGS

Input Supply Voltage, Step-Up Mode . . . . . . . . . . . . . . . 15 V
Input Supply Voltage, Step-Down Mode . . . . . . . . . . . . . 36 V
SW1 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 50 V
SW2 Pin Voltage . . . . . . . . . . . . . . . . . . . . . . . . . 0.5 V to V

Feedback Pin Voltage (ADP1110) . . . . . . . . . . . . . . . . . . 5.5 V
Switch Current . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . . 1.5 A
Maximum Power Dissipation . . . . . . . . . . . . . . . . . . 500 mW
Operating Temperature Range . . . . . . . . . . . . . 0

C to +70

Storage Temperature Range . . . . . . . . . . . . . 65

C to 150

Lead Temperature (Soldering, 10 sec) . . . . . . . . . . . . . 300

TYPICAL APPLICATION

1.5V
AA CELL*

LIM

SW1

SW2

GND

ADP1110-5

47µH

15µF
TANTALUM

SENSE

OPERATES WITH CELL VOLTAGE

1.0V

*ADD 10µF DECOUPLING CAPACITOR IF BATTERY IS

MORE THAN 2' AWAY FROM ADP1110.

Figure 1. 1.5 V to 5 V Converter

ORDERING GUIDE

Model

Output Voltage

Package

ADP1110AN

ADJ

N-8

ADP1110AR

ADJ

SO-8

ADP1110AN-3.3

3.3 V

N-8

ADP1110AR-3.3

3.3 V

SO-8

ADP1110AN-5

5 V

N-8

ADP1110AR-5

5 V

SO-8

ADP1110AN-12

12 V

N-8

ADP1110AR-12

12 V

SO-8

WARNING!

ESD SENSITIVE DEVICE

CAUTION
ESD (electrostatic discharge) sensitive device. Electrostatic charges as high as 4000 V readily
accumulate on the human body and test equipment and can discharge without detection.
Although the ADP1110 features proprietary ESD protection circuitry, permanent damage may
occur on devices subjected to high energy electrostatic discharges. Therefore, proper ESD
precautions are recommended to avoid performance degradation or loss of functionality.

8-Lead SOIC

(SO-8)

LIM

SW1

GND

SW2

FB (SENSE)*

SET

TOP VIEW

(Not to Scale)

ADP1110

*FIXED VERSIONS

JMAX

= 90

= 150

C/W

8-Lead Plastic DIP

(N-8)

LIM

SW1

GND

SW2

*FIXED VERSIONS

FB (SENSE)*

SET

ADP1110

TOP VIEW

(Not to Scale)

JMAX

= 90

= 130

C/W

PIN CONFIGURATIONS

ADP1110-Typical Characteristics

REV. 0

SWITCH

CURRENT A

1.4

0.1

1.4

0.2

0.4

0.5

0.6

0.8

1.2

1.25

1.2

0.6

0.4

0.2

0.8

= +1V

= +1.2V

= +1.5V

= +2V

= +3V

= +5V

SATURATION VOLTAGE V

Figure 2. Saturation Voltage vs. I

SWITCH

Current in Step-Up

Mode

SWITCH

CURRENT A

1.8

0.1

0.9

0.2

0.4

0.6

0.8

1.2

0.6

0.4

0.2

1.6

1.4

0.8

= +12V

ON VOLTAGE V

Figure 3. Switch ON Voltage vs. I

SWITCH

Current In Step-

Down Mode

INPUT VOLTAGE V

1800

1600

1400

1200

1000

800

600

400

200

QUIESCENT CURRENT

QUIESCENT CURRENT µA

Figure 4. Quiescent Current vs. Input Voltage

INPUT VOLTAGE V

OSCILLATOR FREQUENCY kHz

OSCILLATOR FREQUENCY

Figure 5. Oscillator Frequency vs. Input Voltage

LIM

1.9

1.7

0.1

1000

100

1.5

1.3

0.5

1.1

0.9

0.7

0.3

SWITCH CURRENT A

STEP-DOWN WITH
V = +12V

Figure 6. Maximum Switch Current vs. R

LIM

LIM

0.1

1000

100

1.5

1.3

0.5

1.1

0.9

0.7

0.3

STEP-UP MODE
WITH V

+5V

SWITCH CURRENT A

Figure 7. Maximum Switch Current vs. R

LIM

ADP1110

REV. 0

TEMPERATURE C

OSCILLATOR FREQUENCY

OSCILLATOR FREQUENCY kHz

Figure 8. Oscillator Frequency vs. Temperature

TEMPERATURE C

ON TIME µs

9.0

8.9

8.2

8.6

8.5

8.4

8.3

8.8

8.7

SWITCH ON TIME

Figure 9. Switch ON Time vs. Temperature

TEMPERATURE C

DUTY CYCLE

DUTY CYCLE %

Figure 10. Duty Cycle vs. Temperature

TEMPERATURE C

0.54

0.53

0.47

0.51

0.5

0.49

0.48

0.52

= +1.5

SWITCH

= +0.5A

CE(SAT)

 V

Figure 11. Switch ON Voltage Step-Down vs. Temperature

TEMPERATURE C

350

300

200

150

100

250

QUIESCENT CURRENT

QUIESCENT CURRENT µA

Figure 12. Quiescent Current vs. Temperature

TEMPERATURE C

160

120

145

140

135

125

155

150

BIAS CURRENT

BIAS CURRENT nA

130

Figure 13. FB Pin Bias Current vs. Temperature

ADP1110

REV. 0

THEORY OF OPERATION

The ADP1110 is a flexible, low-power, switch-mode power
supply (SMPS) controller. The regulated output voltage can be
greater than the input voltage (boost or step-up mode) or less
than the input (buck or step-down mode). This device uses a
gated-oscillator technique to provide very high performance with
low quiescent current.

A functional block diagram of the ADP1110 is shown on the
first page. The internal 220 mV reference is connected to one
input of the comparator, while the other input is externally
connected (via the FB pin) to a feedback network connected to
the regulated output. When the voltage at the FB pin falls below
220 mV, the 70 kHz oscillator turns on. A driver amplifier provides
base drive to the internal power switch, and the switching action
raises the output voltage. When the voltage at the FB pin exceeds
220 mV, the oscillator is shut off. While the oscillator is off, the
ADP1110 quiescent current is only 300

A. The comparator

includes a small amount of hysteresis, which ensures loop
stability without requiring external components for frequency
compensation.

The maximum current in the internal power switch can be set
by connecting a resistor between V

and the I

LIM

pin. When the

maximum current is exceeded, the switch is turned OFF. The
current limit circuitry has a time delay of about 800 ns. If an
external resistor is not used, connect I

LIM

to V

. Further informa-

tion on I

LIM

is included in the "Applications" section of this data

sheet.

The ADP1110 internal oscillator provides 10

s ON and 5

OFF times, which is ideal for applications where the ratio between
V

and V

OUT

is roughly a factor of three (such as generating +5 V

from a single 1.5 V cell). Wider range conversions, as well as
step-down converters, can also be accomplished with a slight
loss in the maximum output power that can be obtained.

An uncommitted gain block on the ADP1110 can be connected
as a lowbattery detector. The inverting input of the gain block
is internally connected to the 220 mV reference. The noninverting
input is available at the SET pin. A resistor divider, connected
between V

and GND with the junction connected to the SET

pin, causes the AO output to go LOW when the low battery set
point is exceeded. The AO output is an open collector NPN
transistor that can sink 300

The ADP1110 provides external connections for both the
collector and emitter of its internal power switch, which permits

both step-up and step-down modes of operation. For the step-
up mode, the emitter (Pin SW2) is connected to GND and the
collector (Pin SW1) drives the inductor. For step-down mode,
the emitter drives the inductor while the collector is connected
to V

The output voltage of the ADP1110 is set with two external
resistors. Three fixed-voltage models are also available:
ADP11103.3 (+3.3 V), ADP11105 (+5 V) and ADP1110-12
(+12 V). The fixed-voltage models are identical to the
ADP1110 except that laser-trimmed voltage-setting resistors are
included on the chip. Only three external components are
required to form a +3.3 V, +5 V or +12 V converter. On the
fixed-voltage models of the ADP1110, simply connect the
SENSE pin (Pin 8) directly to the output voltage.

COMPONENT SELECTION
General Notes on Inductor Selection

When the ADP1110 internal power switch turns on, current
begins to flow in the inductor. Energy is stored in the inductor
core while the switch is on, and this stored energy is then
transferred to the load when the switch turns off. Because both
the collector and the emitter of the switch transistor are
accessible on the ADP1110, the output voltage can be higher,
lower, or of opposite polarity than the input voltage.

To specify an inductor for the ADP1110, the proper values of
inductance, saturation current, and DC resistance must be
determined. This process is not difficult, and specific equations
for each circuit configuration are provided in this data sheet. In
general terms, however, the inductance value must be low
enough to store the required amount of energy (when both
input voltage and switch ON time are at a minimum) but high
enough that the inductor will not saturate when both V

and

switch ON time are at their maximum values. The inductor
must also store enough energy to supply the load without
saturating. Finally, the dc resistance of the inductor should be
low so that excessive power will not be wasted by heating the
windings. For most ADP1110 applications, an inductor of
15

H to 100

H with a saturation current rating of 300 mA to

1A and dc resistance <0.4

is suitable. Ferrite-core inductors

that meet these specifications are available in small, surface-
mount packages.

To minimize Electro-Magnetic Interference (EMI), a toroid or
pot-core type inductor is recommended. Rod-core inductors are
a lower-cost alternative if EMI is not a problem.

TEMPERATURE C

400

250

200

150

350

300

BIAS CURRENT

BIAS CURRENT nA

100

Figure 14. Set Pin Bias Current vs. Temperature

REF

 mV

TEMPERATURE C

220

219

211

215

214

213

212

217

216

218

REFERENCE VOLTAGE

Figure 15. Reference Voltage vs. Temperature

ADP1110

REV. 0

CALCULATING THE INDUCTOR VALUE

Selecting the proper inductor value is a simple three-step
process:

1. Define the operating parameters: minimum input voltage,

maximum input voltage, output voltage and output current.

2. Select the appropriate conversion topology (step-up, step-

down, or inverting).

3. Calculate the inductor value, using the equations in the

following sections.

INDUCTOR SELECTIONSTEP-UP CONVERTER

In a step-up or boost converter (Figure 19), the inductor must
store enough power to make up the difference between the input
voltage and the output voltage. The power that must be stored
is calculated from the equation:

OUT

IN MIN

(

)

(

)

OUT

(

)

(Equation 1)

where V

is the diode forward voltage (

0.5 V for a 1N5818

Schottky). Because energy is only stored in the inductor while
the ADP1110 switch is ON, the energy stored in the inductor
on each switching cycle must be must be equal to or greater
than:

OSC

(Equation 2)

in order for the ADP1110 to regulate the output voltage.

When the internal power switch turns ON, current flow in the
inductor increases at the rate of:

(t)

R't

(Equation 3)

where L is in Henrys and R' is the sum of the switch equivalent
resistance (typically 0.8

at +25

C) and the dc resistance of

the inductor. If the voltage drop across the switch is small
compared to V

, a simpler equation can be used:

(t)

(Equation 4)

Replacing `t' in the above equation with the ON time of the
ADP1110 (10

s, typical) will define the peak current for a

given inductor value and input voltage. At this point, the
inductor energy can be calculated as follows:

PEAK

(Equation 5)

As previously mentioned, E

must be greater than P

OSC

that the ADP1110 can deliver the necessary power to the load.
For best efficiency, peak current should be limited to 1 A or
less. Higher switch currents will reduce efficiency because of
increased saturation voltage in the switch. High peak current also
increases output ripple. As a general rule, keep peak current as low
as possible to minimize losses in the switch, inductor and diode.

In practice, the inductor value is easily selected using the equations
above. For example, consider a supply that will generate 12 V at
120 mA from a 4.5 V to 8 V source. The inductor power required
is from Equation 1:

12 V

0.5 V

4.5 V

(

)

120 mA

960 mW

On each switching cycle, the inductor must supply:

OSC

960 mW

70 kHz

13.7

Assuming a peak current of 1 A as a starting point, (Equation 4)
can be rearranged to recommend an inductor value:

L(MAX )

4.5V

1 A

Substituting a standard inductor value of 47

H with 0.2

resistance will produce a peak switch current of:

PEAK

4.5V

1.0

862 mA

Once the peak current is known, the inductor energy can be
calculated from Equation 5:

(

)

862 mA

(

)

17.5

Since the inductor energy of 17.5

J is greater than the P

OSC

requirement of 13.7

J, the 47

H inductor will work in this

application. By substituting other inductor values into the same
equations, the optimum inductor value can be determined.
When selecting an inductor, the peak current must not exceed
the maximum switch current of 1.5 A.

The peak current must be evaluated for both minimum and
maximum values of input voltage. If the switch current is high
when V

is at its minimum, the 1.5 A limit may be exceeded at the

maximum value of V

. In this case, the ADP1110's current limit

feature can be used to limit switch current. Simply select a resistor
(using Figure 7) that will limit the maximum switch current to the
I

PEAK

value calculated for the minimum value of V

. This will

improve efficiency by producing a constant I

PEAK

as V

increases.

See the "Limiting the Switch Current" section of this data sheet for
more information.

Note that the switch current limit feature does not protect the
circuit if the output is shorted to ground. In this case, current is
only limited by the dc resistance of the inductor and the forward
voltage of the diode.

INDUCTOR SELECTIONSTEP-DOWN CONVERTER

The step-down mode of operation is shown in Figure 20.

Unlike the step-up mode, the ADP1110's power switch does not
saturate when operating in the step-down mode; therefore,
switch current should be limited to 800 mA in this mode. If the
input voltage will vary over a wide range, the I

LIM

pin can be

used to limit the maximum switch current. Higher switch
current is possible by adding an external switching transistor as
shown in Figure 22.

The first step in selecting the step-down inductor is to calculate
the peak switch current as follows:

PEAK

2 I

OUT

(Equation 6)

where: DC = duty cycle (0.69 for the ADP1110)

= voltage drop across the switch

= diode drop (0.5 V for a 1N5818)

OUT

= output current

OUT

= the output voltage

= the minimum input voltage

ADP1110

REV. 0

As previously mentioned, the switch voltage is higher in step-
down mode than in step-up mode. V

is a function of switch

current and is therefore a function of V

, L, time and V

OUT

For most applications, a V

value of 1.5 V is recommended.

The inductor value can now be calculated:

IN(MIN )

OUT

PEAK

(Equation 7)

where: t

= Switch ON time (10

If the input voltage will vary (such as an application that must
operate from a 9 V, 12 V or 15 V source), an R

LIM

resistor

should be selected from Figure 6. The R

LIM

resistor will keep

switch current constant as the input voltage rises. Note that
there are separate R

LIM

values for step-up and step-down modes

of operation.

For example, assume that +5 V at 250 mA is required from a
+9 V to +18 V source. Deriving the peak current from Equation
6 yields:

PEAK

250 mA

0.69

0.5

1.5

0.5

498 mA

Then, the peak current can be inserted into Equation 7 to
calculate the inductor value:

9 1.5 5

498 mA

Since 50

H is not a standard value, the next lower standard

value of 47

H would be specified.

To avoid exceeding the maximum switch current when the
input voltage is at +18 V, an R

LIM

resistor should be specified.

Using the step-down curve of Figure 6, a value of 560

will

limit the switch current to 500 mA.

INDUCTOR SELECTION--POSITIVE-TO-NEGATIVE
CONVERTER

The configuration for a positive-to-negative converter using the
ADP1110 is shown in Figure 23. As with the step-up converter,
all of the output power for the inverting circuit must be supplied
by the inductor. The required inductor power is derived from
the formula:

P =

OUT

(

)

(

)

(Equation 8)

The ADP1110 power switch does not saturate in positive-to-
negative mode. The voltage drop across the switch can be
modeled as a 0.75 V base-emitter diode in series with a 0.65

resistor. When the switch turns on, inductor current will rise at
a rate determined by:

(t)

R't

(Equation 9)

where: R' = 0.65

+ R

L(DC)

= V

0.75 V

For example, assume that a 5 V output at 75 mA is to be
generated from a +4.5 V to +5.5 V source. The power in the
inductor is calculated from Equation 8:

| 5V|

0.5V

(

)

75 mA

(

)

413 mW

During each switching cycle, the inductor must supply the
following energy:

OSC

413 mW

70 kHz

5.9

Using a standard inductor value of 56

H with 0.2

resistance will produce a peak switch current of:

PEAK

4.5V 0.75V

0.65

0.2

0.85

621 mA

Once the peak current is known, the inductor energy can be
calculated from Equation 9:

(

)

621 mA

(

)

10.8

Since the inductor energy of 10.8

J is greater than the P

OSC

requirement of 5.9

J, the 56

H inductor will work in this

application.

The input voltage only varies between 4.5 V and 5.5 V in this
example. Therefore, the peak current will not change enough to
require an R

LIM

resistor and the I

LIM

pin can be connected

directly to V

. Care should be taken, of course, to ensure that

the peak current does not exceed 800 mA.

CAPACITOR SELECTION

For optimum performance, the ADP1110's output capacitor
must be selected carefully. Choosing an inappropriate capacitor
can result in low efficiency and/or high output ripple.

Ordinary aluminum electrolytic capacitors are inexpensive but
often have poor Equivalent Series Resistance (ESR) and
Equivalent Series Inductance (ESL). Low ESR aluminum
capacitors, specifically designed for switch mode converter
applications, are also available, and these are a better choice
than general purpose devices. Even better performance can be
achieved with tantalum capacitors, although their cost is higher.
Very low values of ESR can be achieved by using OS-CON
capacitors (Sanyo Corporation, San Diego, CA). These devices
are fairly small, available with tape-and-reel packaging and have
very low ESR.

The effects of capacitor selection on output ripple are demon-
strated in Figures 16, 17 and 18. These figures show the output
of the same ADP1110 converter, that was evaluated with three
different output capacitors. In each case, the peak switch
current is 500 mA, and the capacitor value is 100

F. Figure 16

shows a Panasonic HF-series 16-volt radial cap. When the
switch turns off, the output voltage jumps by about 90 mV and
then decays as the inductor discharges into the capacitor. The
rise in voltage indicates an ESR of about 0.18

. In Figure 17,

the aluminum electrolytic has been replaced by a Sprague 293D
series, a 6 V tantalum device. In this case the output jumps

ADP1110

REV. 0

about 30 mV, which indicates an ESR of 0.06

. Figure 18

shows an OS-CON 16volt capacitor in the same circuit, and
ESR is only 0.02

100

1µs

50mV

Figure 16. Aluminum Electrolytic

100

1µs

50mV

Figure 17. Tantalum Electrolytic

100

1µs

50mV

Figure 18. OS-CON Capacitor

If low output ripple is important, the user should consider the
ADP3000. Because this device switches at 400 kHz, lower peak
current can be used. Also, the higher switching frequency
simplifies the design of the output filter. Consult the ADP3000
data sheet for additional details.

DIODE SELECTION

In specifying a diode, consideration must be given to speed,
forward voltage drop and reverse leakage current. When the
ADP1110 switch turns off, the diode must turn on rapidly if
high efficiency is to be maintained. Shottky rectifiers, as well as
fast signal diodes such as the 1N4148, are appropriate. The
forward voltage of the diode represents power that is not
delivered to the load, so V

must also be minimized. Again,

Schottky diodes are recommended. Leakage current is especially

important in low-current applications, where the leakage can be
a significant percentage of the total quiescent current.

For most circuits, the 1N5818 is a suitable companion to the
ADP1110. This diode has a V

of 0.5 V at 1 A, 4

A to 10

leakage, and fast turn-on and turn-off times. A surface mount
version, the MBRS130LT3, is also available.

For switch currents of 100 mA or less, a Shottky diode such as
the BAT85 provides a V

of 0.8 V at 100 mA and leakage less

than 1

A. A similar device, the BAT54, is available in a SOT23

package. Even lower leakage, in the 1 nA to 5 nA range, can be
obtained with a 1N4148 signal diode.

General purpose rectifiers, such as the 1N4001, are not suitable
for ADP1110 circuits. These devices, which have turn-on times
of 10

s or more, are too slow for switching power supply

applications. Using such a diode "just to get started" will result
in wasted time and effort. Even if an ADP1110 circuit appears
to function with a 1N4001, the resulting performance will not
be indicative of the circuit performance when the correct diode
is used.

CIRCUIT OPERATION, STEP-UP (BOOST) MODE

In boost mode, the ADP1110 produces an output voltage that is
higher than the input voltage. For example, +5 V can be derived
from one alkaline cell (+1.5 V), or +12 V can be generated from
a +5 V logic power supply.

Figure 19 shows an ADP1110 configured for step-up operation.
The collector of the internal power switch is connected to the
output side of the inductor, while the emitter is connected to
GND. When the switch turns on, pin SW1 is pulled near
ground. This action forces a voltage across L1 equal to V

CE(SAT),

and current begins to flow through L1. This current

reaches a final value (ignoring second-order effects) of:

PEAK

CE(SAT )

where 10

s is the ADP1110 switch's "on" time.

LIM

SW1

SW2

GND

ADP1110

OUT

R3*

*OPTIONAL

Figure 19. Step-Up Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, current begins to flow
through D1 into the load, and the output voltage is driven above
the input voltage.

The output voltage is fed back to the ADP1110 via resistors R1
and R2. When the voltage at pin FB falls below 220 mV, SW1

ADP1110

REV. 0

turns "on" again, and the cycle repeats. The output voltage is
therefore set by the formula:

OUT

220 mV

The circuit of Figure 19 shows a direct current path from V

OUT

, via the inductor and D1. Therefore, the boost converter

is not protected if the output is short circuited to ground.

CIRCUIT OPERATION, STEP-DOWN (BUCK) MODE

The ADP1110's step-down mode is used to produce an output
voltage that is lower than the input voltage. For example, the
output of four NiCd cells (+4.8 V) can be converted to a +3 V
logic supply.

A typical configuration for step-down operation of the ADP1110 is
shown in Figure 20. In this case, the collector of the internal
power switch is connected to V

and the emitter drives the

inductor. When the switch turns on, SW2 is pulled up towards
V

. This forces a voltage across L1 equal to V

OUT

and causes current to flow in L1. This current reaches a final
value of:

PEAK

OUT

where 10

s is the ADP1110 switch's "on" time.

LIM

SW1

SW2

GND

SET

ADP1110

D1
1N5818

OUT

LIM

100

Figure 20. Step-Down Mode Operation

When the switch turns off, the magnetic field collapses. The
polarity across the inductor changes, and the switch side of the
inductor is driven below ground. Schottky diode D1 then turns
on, and current flows into the load. Notice that the Absolute
Maximum Rating for the ADP1110's SW2 pin is 0.5 V below
ground. To avoid exceeding this limit, D1 must be a Schottky
diode. Using a silicon diode in this application will generate
forward voltages above 0.5 V that will cause potentially
damaging power dissipation within the ADP1110.

The output voltage of the buck regulator is fed back to the
ADP1110's FB pin by resistors R1 and R2. When the voltage at
pin FB falls below 220 mV, the internal power switch turns
"on" again and the cycle repeats. The output voltage is set by
the formula:

OUT

220 mV

When operating the ADP1110 in step-down mode, the output
voltage is impressed across the internal power switch's emitter-

base junction when the switch is off. To protect the switch, the
output voltage should be limited to 6.2 V or less. If a higher
output voltage is required, a Schottky diode should be placed in
series with SW2, as shown in Figure 21.

LIM

SW1

SW2

GND

SET

ADP1110

D1
1N5818

OUTPUT

INPUT

LIM

Figure 21. Step-Down Mode, V

OUT

> 6.2 V

If the input voltage to the ADP1110 varies over a wide range, a
current limiting resistor at Pin 1 may be required. If a particular
circuit requires high peak inductor current with minimum input
supply voltage, the peak current may exceed the switch maxi-
mum rating and/or saturate the inductor when the supply
voltage is at the maximum value. See the "Limiting the Switch
Current" section of this data sheet for specific recommendations.

INCREASING OUTPUT CURRENT IN THE STEP-DOWN
REGULATOR

Unlike the boost configuration, the ADP1110's internal power
switch is not saturated when operating in step-down mode. A
conservative value for the voltage across the switch in step-down
mode is 1.5 V. This results in high power dissipation within the
ADP1110 when high peak current is required. To increase the
output current, an external PNP switch can be added (Figure
22). In this circuit, the ADP1110 provides base drive to Q1
through R3, while R4 ensures that Q1 turns off rapidly. Because
the ADP1110's internal current limiting function will not work
in this circuit, R5 is provided for this purpose. With the value
shown, R5 limits current to 2 A. In addition to reducing power
dissipation on the ADP1110, this circuit also reduces the switch
voltage. When selecting an inductor value for the circuit of
Figure 22, the switch voltage can be calculated from the
formula:

Q1(SAT )

0.6V

0.4V

LIM

SW1

SW2

GND

SET

ADP1110

1N5821

OUTPUT

INPUT

LIM

INPUT

220

330

MJE210

0.3

Figure 22. High Current Step-Down Operation

ADP1110

REV. 0

POSITIVE-TO-NEGATIVE CONVERSION

The ADP1110 can convert a positive input voltage to a negative
output voltage as shown in Figure 23. This circuit is essentially
identical to the step-down application of Figure 19, except that
the "output" side of the inductor is connected to power ground.
When the ADP1110's internal power switch turns off, current
flowing in the inductor forces the output (V

OUT

) to a negative

potential. The ADP1110 will continue to turn the switch on
until its FB pin is 220 mV above its GND pin, so the output
voltage is determined by the formula:

OUT

220 mV

ILIM

VIN

SW1
SW2

GND

SET

ADP1110

1N5818

OUTPUT

INPUT

LIM

INPUT

NEGATIVE
OUTPUT

Figure 23. A Positive-to-Negative Converter

The design criteria for the step-down application also apply to
the positive-to-negative converter. The output voltage should be
limited to |6.2 V| unless a diode is inserted in series with the
SW2 pin (see Figure 21.) Also, D1 must again be a Schottky
diode to prevent excessive power dissipation in the ADP1110.

NEGATIVE-TO-POSITIVE CONVERSION

The circuit of Figure 24 converts a negative input voltage to a
positive output voltage. Operation of this circuit configuration is
similar to the step-up topology of Figure 19, except the current
through feedback resistor R1 is level-shifted below ground by a
PNP transistor. The voltage across R1 is V

OUT

BEQ1

. However,

diode D2 level-shifts the base of Q1 about 0.6 V below ground
thereby cancelling the V

of Q1. The addition of D2 also reduces

the circuit's output voltage sensitivity to temperature, which other-
wise would be dominated by the 2 mV V

contribution of Q1.

The output voltage for this circuit is determined by the formula:

OUT

220 mV

Unlike the positive step-up converter, the negative-to-positive
converter's output voltage can be either higher or lower than the
input voltage.

LIM

SW1

SW2

GND

SET

ADP1110

INPUT

LIM

POSITIVE
OUTPUT

10K

NEGATIVE

INPUT

2N3906

1N4148

Figure 24. A Negative-to-Positive Converter

LIMITING THE SWITCH CURRENT

The ADP1110's R

LIM

pin permits the switch current to be

limited with a single resistor. This current limiting action occurs
on a pulse by pulse basis. This feature allows the input voltage
to vary over a wide range without saturating the inductor or
exceeding the maximum switch rating. For example, a particular
design may require peak switch current of 800 mA with a 2.0 V
input. If V

rises to 4 V, however, the switch current will

exceed 1.6 A. The ADP1110 limits switch current to 1.5 A and
thereby protects the switch, but the output ripple will increase.
Selecting the proper resistor will limit the switch current to
800 mA, even if V

increases. The relationship between R

LIM

and maximum switch current is shown in Figure 6.

The I

LIM

feature is also valuable for controlling inductor current

when the ADP1110 goes into continuous-conduction mode.
This occurs in the step-up mode when the following condition is
met:

OUT

D IODE

1 DC

where DC is the ADP1110's duty cycle. When this relationship
exists, the inductor current does not go all the way to zero
during the time that the switch is OFF. When the switch turns
on for the next cycle, the inductor current begins to ramp up
from the residual level. If the switch ON time remains constant,
the inductor current will increase to a high level (see Figure 25).
This increases output ripple and can require a larger inductor
and capacitor. By controlling switch current with the I

LIM

resistor, output ripple current can be maintained at the design
values. Figure 26 illustrates the action of the I

LIM

circuit.

100

10µs

10mV

200mA/div.

Figure 25. I

LIM

Operation--I

Characteristic

100

10µs

10mV

200mA/div.

Figure 26. I

LIM

Operation--I

Characteristic

ADP1110

REV. 0

The internal structure of the I

LIM

circuit is shown in Figure 27.

Q1 is the ADP1110's internal power switch, that is paralleled by
sense transistor Q2. The relative sizes of Q1 and Q2 are scaled
so that I

is 0.5% of I

. Current flows to Q2 through an

internal 80

resistor and through the R

LIM

resistor. These two

resistors parallel the base-emitter junction of the oscillator-
disable transistor, Q3. When the voltage across R1 and R

LIM

exceeds 0.6 V, Q3 turns on and terminates the output pulse. If
only the 80

internal resistor is used (i.e., the I

LIM

pin is

connected directly to V

), the maximum switch current will be

1.5 A. Figure 6 gives R

LIM

values for lower current-limit values.

72kHz
OSC

POWER
SWITCH

SW2

SW1

LIM

DRIVER

(INTERNAL)

LIM

IQ1

200

(EXTERNAL)

ADP1110

Figure 27. ADP1110 Current Limit Operation

The delay through the current limiting circuit is approximately
800 ns. If the switch ON time is reduced to less than 3

accuracy of the current trip-point is reduced. Attempting to
program a switch ON time of 800 ns or less will produce
spurious responses in the switch ON time; however, the
ADP1110 will still provide a properly-regulated output voltage.

PROGRAMMING THE GAIN BLOCK

The gain block of the ADP1110 can be used as a low-battery
detector, error amplifier or linear post regulator. The gain block
consists of an op amp with PNP inputs and an open-collector
NPN output. The inverting input is internally connected to the
ADP1110's 220 mV reference, while the noninverting input is
available at the SET pin. The NPN output transistor will sink
about 300

Figure 28 shows the gain block configured as a low-battery
monitor. Resistors R1 and R2 should be set to high values to
reduce quiescent current, but not so high that bias current in
the SET input causes large errors. A value of 33 k

for R2 is a

good compromise. The value for R1 is then calculated from the
formula:

LOBATT

220 mV

where V

LOBATT

is the desired low battery trip point. Since the

gain block output is an open-collector NPN, a pull-up resistor
should be connected to the positive logic power supply.

ADP1110

GND

BAT

SET

33k

220V
V

REF

HYS

LOGIC

Figure 28. Setting the Low Battery Detector Trip Point

The circuit of Figure 28 may produce multiple pulses when
approaching the trip point due to noise coupled into the SET
input. To prevent multiple interrupts to the digital logic,
hysteresis can be added to the circuit. Resistor R

HYS

, with a

value of 1 M

to 10 M

, provides the hysteresis. The addition

of R

HYS

will change the trip point slightly, so the new value for

R1 will be:

LOBATT

220 mV

HYS

where V

is the logic power supply voltage, R

is the pull-up

resistor, and R

HYS

creates the hysteresis.

The gain block can also be used as a control element to reduce
output ripple. The ADP3000 is normally recommended for low-
ripple applications, but its minimum input voltage is 2 V. The
gain-block technique using the ADP1110 can be useful for step-
up converters operating down to 1 V.

A step-up converter using this technique is shown in Figure 29.
This configuration uses the gain block to sense the output
voltage and control the comparator. The result is that the
comparator hysteresis is reduced by the open loop gain of the
gain block. Output ripple can be reduced to only a few millivolts
with this technique, versus a typical value of 90 mV for a +5 V
converter using just the comparator. For best results, a large
output capacitor (1000

F or more) should be specified. This

technique can also be used for step-down or inverting applica-
tions, but the ADP3000 is usually a more appropriate choice.
See the ADP3000 data sheet for further details.

LIM

SW1

SW2

FB GND

SET

ADP1110

OUTPUT

INPUT

1000µF

300k

13.8k

270k

10µF

CTX15-4

1N5818

15µH

Figure 29. Using the Gain Block to Reduce Output Ripple

ADP1110

REV. 0

APPLICATION CIRCUITS
All-Surface-Mount, Single-Cell to 5 V Converter

This is a very simple, compact, low-part-count circuit that takes
a single alkaline 1.5 V cell input and produces a 5 V output.
The output current should be kept to 10 mA or less to conserve
battery life.

LIM

SW1

SW2

SENSE

GND

SET

ADP1110-5

1N5817

50µH

+5V
10mA

15µF

ONE

ALKALINE

CELL

CTX50-4

1.5V

Figure 30. All-Surface-Mount, Single-Cell to 5 V Converter

All-Surface-Mount, 3 V to 5 V Step-Up Converter

Similar to the previous circuit, this circuit takes a 3-volt input
and provides a 5 V output at 40 mA. As in the single-cell version,
the circuit is compact and uses only four external components.

LIM

SW1

SW2

SENSE

GND

SET

ADP1110-5

220

50µH

+5V
40mA

10µF

TWO

ALKALINE

CELLS

1N5817

CTX50-4

Figure 31. All-Surface-Mount, 3 V to 5 V Step-Up
Converter

All-Surface-Mount, 9 V to 5 V Step-Down Converter

Featuring the same low parts count of the step-up design, this
circuit is the complement to the preceding one. The 220

resistor programs the current limit to around 600 mA.

LIM

SW1
SW2

SENSE

GND

SET

ADP1110-5

220

50µH

1N5817

+5V
40mA

10µF

BATTERY

LIM

CTX50-4

Figure 32. All-Surface-Mount, 9 V to 5 V Step-Down
Converter

1.5 V to 5 V Dual-Output Step-Up Converter

This circuit works from a single 1.5 V cell and provides simulta-
neous outputs of +5 V and 5 V. The accuracy of the negative
output suffers slightly because of the extra diode drop of around
0.4 V.

LIM

SW1

SW2

SENSE

GND

SET

ADP1110-5

1.5V

L16
8µH

4.7µF

ONE

ALKALINE

CELL

4.7µF

+5V
3mA

NOTE: ALL DIODES 1N5818

5V
3mA

4.7µF

CTX68-4

Figure 33. 1.5 V to

5 V Dual-Output Step-Up

Converter

All-Surface-Mount Flash Memory VPP Generator

Figure 34 shows a circuit that can generate the programming
voltage, VPP to program flash memory. The key components
are the MOSFET and the bipolar transistor. These two devices
form a switch that, when ON, allows the ADP1110 to power-up
and function as a step-up converter. The output is +12 V at 120
mA. When the MOSFET switch is OFF, the output of the
circuit drops to just under +5 V thereby disabling the program-
ming capability.

Care should be taken so there is no short-circuit-current limiting
in the circuit in either operating mode.

LIM

SW1

SW2

SENSE

GND

SET

ADP1110-12

50µH

10µF

LOGIC1 = PROGRAM
LOGIC0 = SHUTDOWN

1N5818

CTX50-4

MMBT4403

MMBF170

+12V
120mA

+5V

10k

Figure 34. All Surface-Mount Flash Memory VPP Generator

ADP1110

REV. 0

1.5 V to +5 V, +10 V Dual Output Step-Up Converter

The circuit of Figure 35 illustrates a way to get outputs of
+10 V and +5 V from the same converter. The main 5 V output
is derived from the feedback provided by the 487 k

and 11 k

resistors. Capacitor C1 should be a multilayer ceramic variety
for best performance, but a good quality tantalum capacitor will
also give good performance at lower cost.

LIM

SW1

SW2

GND

SET

ADP1110

1.5V

50µH

4.7µF

ONE

ALKALINE

CELL

4.7µF

NOTE: ALL DIODES 1N5818

CTX50-4

+5V
3mA

11k

487k

+10V
3mA

220

Figure 35. 1.5 V to +5 V, +10 V Dual Output Step-Up
Converter

1.5 V-Powered Laser Diode Driver

Figure 36 shows a circuit suitable for driving many laser diodes
that incorporate a photodiode to monitor the laser diode
current, this circuit makes use of the gain block and current-
limit functions to provide a feedback system based on the
average laser diode current. This current must be controlled
very closely or permanent damage to the laser diode is likely to
be the result.

To ensure that the laser is operating at the proper power level,
the actual optical power from the laser should be monitored
with a calibrated photodiode or optical power meter. In
addition, the actual diode current should also be monitored, and
R1 can be adjusted to give the correct output power.

NOTES
1. All inductors referenced are Coiltronics CTX-series except

where noted.

2. If the source of power is more than an inch or so from the

converter, the input to the converter should be bypassed with
approximately 10

F of capacitance. This capacitor should be

a good quality tantalum or aluminum electrolytic.

LIM

SW1

SW2

GND

SET

ADP1110

1N4148

220

1.5V

5.1k

0.022µF

2.2µH

1µF

1N5818

MJE210

TOSHIBA
TOLD-9321

100µF

OS-Con

2N3906

Figure 36. 1.5 V-Powered Laser Diode Driver

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